Active clamp flyback converters and control methods thereof

ABSTRACT

A control method for a power convert is disclosed. The power convert uses an active-clamp flyback topology and has low-side and high-side switches. The low-side switch is switched to generate consecutive switching cycles including a modified flyback cycle and a normal flyback cycle. Each of the consecutive switching cycles is not less than a blanking time generated in response to a load of the power converter. The high-side switch is constantly turned OFF during the normal flyback cycle. The high-side switch is turned ON after the blanking time during the modified flyback cycle to perform zero-voltage switching for the low-side switch.

CROSS-REFERENCE TO RELATED APPLICATION

This application claims priority to and the benefit of Taiwan Application Series Number 107134308 filed on Sep. 28, 2018, which is incorporated by reference in its entirety.

BACKGROUND

The present disclosure relates generally to power converters using an active-clamp flyback topology and control methods thereof, and, more particularly, to control methods that operates a power converter using an active-clamp flyback topology in one of several operation modes.

Flyback power converters have been widely adopted in the power supplies of home appliances, computers, battery chargers for example. To further improve the efficiency of a flyback power converter, an active-clamp topology has been introduced, using an active-clamp circuit to replace a snubber, which is commonly used to consume the energy stored by the leakage inductance of a primary winding of a transformer in the flyback power converter. A power converter using an active-clamp flyback topology is named an ACF power converter in short. ACF power converter is well known to have outstanding power efficiency when a load of the ACF power converter is heavy. However, the power efficiency deteriorates seriously when the load is light, substantially due to the significant circulated current continuously going back and forth through a primary winding of the ACF power converter.

Texas Instruments introduces UCC28780, a controller used in an ACF power converter. UCC28780 is capable of operating in one of four operation modes, basically in response to the state of the load of the ACF power converter. The circuit application datasheet of USCC28780, however, still shows a bleeder resistor connected in parallel with a capacitor of an active-clamp circuit, to slowly release the energy accumulated on the capacitor. Obviously, as the bleeder resistor acts as an energy eater, USCC28780 does completely employ the benefit of the active-clamp circuit.

Furthermore, system designers of conventional ACF power converters usually confront the difficulties in dealing with electromagnetic interference (EMI) and audible noise.

BRIEF DESCRIPTION OF THE DRAWINGS

Non-limiting and non-exhaustive embodiments of the present invention are described with reference to the following drawings. In the drawings, like reference numerals refer to like parts throughout the various figures unless otherwise specified. These drawings are not necessarily drawn to scale. Likewise, the relative sizes of elements illustrated by the drawings may differ from the relative sizes depicted.

The invention can be more fully understood by the subsequent detailed description and examples with references made to the accompanying drawings, wherein:

FIG. 1 shows ACF power converter 10 according embodiments of the invention;

FIG. 2 demonstrates two operation modes, one called hereinafter the ACF mode and the other the flyback mode;

FIG. 3A shows some signal waveforms when ACF power converter 10 is operated in the ACF mode;

FIG. 3B shows some signal waveforms when ACF power converter 10 is operated in the flyback mode;

FIG. 4 enlarges the waveform of current-sense signal V_(CS) during low-side ON time T_(ON-L);

FIG. 5A shows the relationship between switching frequency f_(CYC) and compensation signal V_(COMP);

FIG. 5B demonstrates the relationship between signal peak V_(CS-PEAK) and compensation signal V_(COMP) regarding to the embodiment of FIG. 1;

FIG. 5C demonstrates the graph of current I_(O) vs compensation signal V_(COMP) when the embodiment of FIG. 1 stays on a steady state, and the mode switching between the ACF mode and the flyback mode as well;

FIG. 6 demonstrates consecutive switching cycles T_(CYC) when ACF power converter 10 in FIG. 1 is operated in the flyback mode; and

FIG. 7 demonstrates control method 60 in use of power controller 14.

DETAILED DESCRIPTION

FIG. 1 shows ACF power converter 10 according embodiments of the invention. Bridge rectifier BD performs full-wave rectification for alternating-current voltage V_(AC) from a power grid, to provide input power lines IN and GNDI. Input voltage V_(IN) at input power line IN is positive in reference to the voltage at input power line GNDI, which is referred to as input ground GNDI hereinafter. Transformer TF includes primary winding LP, secondary winding LS and auxiliary winding LA, inductively coupled to each other. Primary wilding LP, low-side switch LSS, and current-sense resistor RCS are connected in series between input power line IN and input ground GNDI. Low-side switch LSS and current-sense resistor connect primary wilding LP to input ground GNDI. Current-sense resistor RCS provides current-sense signal V_(CS) to power controller 14, via current-sense pin CS. High-side switch HSS connects in series with capacitor CAC to form active-clamp circuit ACC, which connects in parallel with primary winding LP. When low-side switch LSS is turned ON, current-sense signal V_(CS) is representative of inductor current I_(M) flowing through primary winding LP.

Power controller 14, an integrated circuit in an embodiment, controls driver DVR, which could be another integrated circuit in one embodiment of the invention, to provide to high-side switch HSS and low-side switch LSS high-side signal DRV_(HS) and low-side signal DRV_(LS) respectively. High-side switch HSS and low-side switch LSS could be high-voltage GaN transistors or MOS transistors in some embodiments of the invention. In some embodiments of the invention, driver DVR, high-side switch HSS and low-side switch, individually manufactured, are all integrated into one integrated-circuit package. Power controller 14 and driver DVR, in combination, could act as a control circuit to provide high-side signal DRV_(HS) and low-side signal DRV_(HS), controlling high-side switch HSS and low-side switch LSS respectively.

By turning ON and OFF high-side switch HSS and low-side switch LSS, power controller 14 causes inductor current I_(M) to vary, so that secondary winding LS reflectively generates alternating-current voltage, which is then rectified to provide output power lines OUT and GNDO. Output voltage V_(OUT) at output power line OUT is positive in reference to the voltage at output power line GNDO, which is referred to as output ground GNDO hereinafter. Output voltage V_(OUT) provides output current I_(O) to charge or power load 13, a rechargeable battery for instance.

To regulate output voltage V_(OUT), power controller 14 acquires negative feedback control from the combination of error amplifier EA, optical coupler OPT and compensation capacitor CCOMP. At the secondary side, error amplifier EA compares output voltage V_(OUT) with target voltage V_(REF-TAR), to control compensation signal V_(COMP) on compensation capacitor CCOMP at the primary side, via optical coupler OPT that provides DC isolation between the primary and secondary sides. For example, when output voltage V_(OUT) exceeds target voltage V_(REF-TAR), compensation signal V_(COMP) goes down and the power that ACF power converter 10 converts to load 13 reduces, so as to regulate output voltage V_(OUT) about at target voltage V_(REF-TAR).

AC voltage is induced across auxiliary winding LA at the primary side, and rectified to generate operating voltage V_(CC) at power input pin VCC of power controller 14, where operating voltage V_(CC) substantially supplies the power needed for the operation of power controller 14. Resisters RA and RB, connected in series, form a voltage-dividing circuit, and the joint between resisters RA and RB is connected to feedback pin FB of power controller 14. Feedback voltage V_(FB) is at feedback pin FB.

Power controller 14 and driver DVR generate high-side signal DRV_(HS) and low-side signal DRV_(LS) in response to current-sense signal V_(CS), compensation signal V_(COMP) and feedback voltage V_(FB).

In one embodiment of the invention, power controller 14 adaptively switches to operate in one of two operation modes, but the invention is not limited to, however. In another embodiment of the invention, power controller 14 adaptively switches to operate in one of three operation modes. FIG. 2 demonstrates two operation modes, one called hereinafter the ACF mode and the other the flyback mode. Generally speaking, the ACF mode is used when load 13 is in a heavy state, and the flyback mode is used when load 13 is in a light state or there is no load.

As demonstrated in FIG. 2, when operated in the ACF mode, power controller 14 is configured to perform: 1) making high-side signal DRV_(HS) and low-side signal DRV_(LS) substantially complementary to each other, and high-side switch HSS and low-side switch LSS perform ZVS; 2) fixing but jittering switching frequency f_(CYC); and 3) modulating signal peak V_(CS-PEAK) in response to compensation signal V_(COMP), where signal peak V_(CS-PEAK) is a local maximum of current-sense signal V_(CS) and will be detailed later. Power controller 14, when operated in the ACF mode, checks whether positive-current duration T_(ON-P) and negative-current duration T_(ON-N) fit a predetermined relationship to exit the ACF mode and enter the flyback mode. Positive-current duration T_(ON-P) and negative-current duration T_(ON-N) represent durations when current-sense signal V_(CS) is positive and negative respectively, especially when low-side switch LSS is turned ON.

In comparison with compensation signal V_(COMP), a predetermined relationship between positive-current duration T_(ON-P) and negative-current duration T_(ON-N) could be more suitable to indicate the state of load 13 when power controller 14 is operated in the ACF mode, and could be used as an indicator to switch operation modes.

When operated in the flyback mode, power controller 14 is configured to perform: 1) keeping high-side switch HSS substantially turned OFF; 2) making signal peak V_(CS-PEAK) about a constant; and 3) modulating and jittering switching frequency f_(CYC) in response to compensation V_(COMP) Meanwhile, power controller 14 monitors whether the compensation signal V_(COMP) exceeds a reference voltage V_(COMP-REF), to exit the flyback mode and enter the ACF mode.

Please refer to FIGS. 2 and 3A, where FIG. 3A shows some signal waveforms when ACF power converter 10 is operated in the ACF mode. From top to bottom, signal waveforms in FIG. 3A are clock signal CLK internally generated by power controller 14, high-side signal DRV_(HS), low-side signal DRV_(LS), current-sense signal V_(CS), terminal voltage V_(SW) at the joint between high-side switch HSS and low-side switch LSS, and winding voltage V_(AUX) across auxiliary winding LA.

Power controller 14 has in itself a clock generator to provide clock signal CLK, capable of defining switching cycle T_(CYC), the reciprocal of which is switching frequency f_(CYC) of low-side signal DRV_(LS).

When power controller 14 is operated in the ACF mode, switching frequency f_(CYC) is about a constant independent from compensation signal V_(COMP), and might optionally be jittered. For example, during the time when power controller 14 is operated in the ACF mode, switching frequency f_(CYC) is independent from compensation signal V_(COMP), centers at 200 kHz and varies periodically between 190 kHz and 210 kHz with a jittering frequency of 400 Hz, capable of solving electromagnetic interference (EMI) issues.

When operated in the ACF mode, power controller 14 makes high-side signal DRV_(HS) and low-side signal DRV_(LS) substantially complementary, as demonstrated by the waveforms in FIG. 3A. The ACF mode is a complementary mode, therefore. When high-side signal DRV_(HS) turns from logic “1” to logic “0”, dead time TD_(F) follows and then low-side signal DRV_(LS) complementarily turns from logic “0” to logic “1”. Similarly, when low-side signal DRV_(LS) turns from logic “1” to logic “0”, dead time TD_(R) follows and then high-side signal DRV_(HS) complementarily turns from logic “0” to logic “1”.

Dead times TD_(F) and TD_(R) are short but necessary. Their existence prevents the short through happening when both high-side switch HSS and low-side switch LSS are turned ON at the same time, and also helps high-side switch HSS and low-side switch LSS both to perform zero-voltage switching (ZVS). It is known in the art that low-side signal DRV_(LS) and high-side signal DRV_(HS) are substantially complementary to each other even though they both are “0” in logic briefly during dead times TD_(F) and TD_(R). For example, when low-side signal DRV_(LS) turns from logic “1” into logic “0”, winding voltage V_(AUX) raises from a negative voltage V_(N) and approaches to a positive voltage V_(P), while terminal voltage V_(SW) raises from 0V to approach voltage V_(CP), as shown in FIG. 3A. Voltage V_(CP) is the voltage at the joint between high-side switch HSS and capacitor CAC. Power controller 14 senses winding voltage V_(AUX) by detecting feedback voltage V_(FB). Once it is found that winding voltage V_(AUX) is about the positive voltage V_(P), it can be determined that terminal voltage V_(SW) is about voltage V_(CP), and accordingly power controller 14 changes high-side signal DRV_(HS) from “0” into “1” in logic, performing ZVS at high-side switch HSS. Similarly, when high-side signal DRV_(HS) turns from logic “1” into logic “0”, power controller 14 could detect winding voltage V_(AUX) to know whether terminal voltage V_(SW) drops to be about 0V, and when it is determined that the terminal voltage V_(SW) is about 0V, changes low-side signal DRV_(LS) from “0” into “1” in logic, performing ZVS at low-side switch LSS.

Low-side ON time T_(ON-L) refers to the period of time when low-side signal DRV_(LS) is “1” in logic, or the period of time when low-side switch LSS conducts current. Analogously, high-side ON time T_(ON-H) is the period of time when high-side signal DRV_(HS) is “1” in logic, or the period of time when high-side switch HSS conducts current.

FIG. 3A also shows how power controller 14 modulates signal peak V_(CS-PEAK). In FIG. 3A attenuated compensation signal V_(COMP-SC) is in a linear correlation with compensation signal V_(COMP). For example, V_(COMP-SC)=K*V_(COMP), where K is a constant between 0 and 1. A voltage divider comprising resistors connected in series, for example, divides compensation signal V_(COMP) to generate attenuated compensation signal V_(COMP-SC). Attenuated compensation signal V_(COMP-SC) controls signal peak V_(CS-PEAK). During low-side ON time T_(ON-L), current-sense signal V_(CS) increases over time, and when determining that current-sense signal V_(CS) exceeds attenuated compensation signal V_(COMP-SC), power controller 14 ends low-side ON time T_(ON-L) and, further after a delay of dead time TD_(R), starts high-side ON time T_(ON-H). During dead time TD_(R), current-sense signal V_(CS) drops, and the local maximum of current-sense signal V_(CS) becomes signal peak V_(CS-PEAK), which is about attenuated compensation signal V_(COMP-SC), as shown in FIG. 3A. Accordingly, power controller 14 modulates signal peak V_(CS-PEAK) in response to compensation signal V_(COMP) In comparison with the switching cycle at the left portion of FIG. 3A, the one at the right portion or FIG. 3A has a larger attenuated compensation V_(COMP-SC), so signal peak V_(CS-PEAK) is larger in the right portion or FIG. 3A. In other words, power controller 14 makes signal peak V_(CS-PEAK) in a linear correlation with compensation signal V_(COMP).

A switching cycle T_(CYC) shown in FIG. 3A consists of dead time TD_(F), low-side ON time T_(ON-L), dead time TD_(R) and high-side ON time T_(ON-H). A pulse of clock signal CLK ends high-side ON time T_(ON-H) and starts dead time TD_(F), which ends at about the moment when terminal voltage V_(SW) is 0V. Low-side ON time T_(ON-L) follows dead time TD_(F), and ends when current-sense signal V_(CS) exceeds attenuated compensation signal V_(COMP-SC) Dead time TD_(R) follows low-side ON time T_(ON-L), and ends when terminal voltage V_(SW) is about voltage V_(CP), to start high-side ON time T_(ON-H). A next pulse of clock signal CLK ends high-side ON time T_(ON-H) and also concludes a switching cycle T_(CYC).

When operated at the ACF mode, inductor current I_(M) flowing through primary winding LP does not stop at 0 A, always changing.

Please refer to FIGS. 2 and 3B, where FIG. 3B shows some signal waveforms when ACF power converter 10 is operated in the flyback mode. From top to bottom, signal waveforms in FIG. 3B are clock signal CLK, high-side signal DRV_(HS), low-side signal DRV_(HS), current-sense signal V_(CS), terminal voltage V_(SW), and winding voltage V_(AUX).

As shown in FIG. 3B, when operated in the flyback mode, high-side signal DRV_(HS) is substantially kept as “0” in logic to turn high-side switch HSS OFF, and low-side signal DRV_(LS) periodically switches low-side switch LSS. The flyback mode is a non-complementary mode because high-side signal DRV_(HS) and low-side signal DRV_(LS) are not complementary to each other, obviously.

Shown in FIG. 3B, a pulse of clock signal CLK starts a switching cycle T_(CYC) and low-side ON time T_(ON-L) as well. When current-sense signal V_(CS) exceeds a constant reference voltage V_(CS-REF), low-side ON time T_(ON-L) ends and demagnetization time T_(DMG) starts. Reference voltage V_(CS-REF) is independent to compensation signal V_(COMP) During demagnetization time T_(DMG), secondary winding LS releases energy to build up output voltage V_(OUT). Demagnetization time T_(DMG) comes to an end when secondary winding LS completely depletes the energy it carries, so terminal voltage V_(SW) starts oscillating, and oscillation time T_(OSC) begins, as shown in FIG. 3B. A next pulse of clock signal CLK concludes both oscillation time T_(OSC) and a switching cycle T_(CYC). When operated in the flyback cycle, a switching cycle time T_(CYC) consists of low-side ON time T_(ON-L), demagnetization time T_(DMG) and oscillation time T_(OSC).

In FIG. 3B, when operated in the flyback mode, peak signal V_(CS-PEAK) is independent to the variation of attenuated compensation signal V_(COMP-SC) or compensation signal V_(COMP), and is about a constant substantially equal to reference voltage V_(CS-REF).

When operated in the flyback mode, the clock generator providing clock signal CLK is controlled by compensation signal V_(COMP). By comparing the left and the right portions of FIG. 3B, it can be found that the lower attenuated compensation signal V_(COMP-SC) the longer switching cycle T_(CYC).

When power controller 14 is operated in the flyback mode, switching frequency f_(CYC), the reciprocal of switching cycle T_(CYC), depends on compensation signal V_(COMP), and might optionally be jittered to solve EMI issues. For example, during the time when power controller 14 is operated in the flyback mode, switching frequency f_(CYC) centers at an average frequency and varies periodically between upper and lower frequencies, where the average frequency is a function of compensation signal V_(COMP).

Even though FIG. 3B shows that high-side switch HSS is constantly turned OFF, the invention is not limited to however. In another embodiment of the invention, when power controller 14 is operated in the flyback mode, high-side switch HSS is not turned ON during low-side ON time T_(ON_L) and demagnetization time T_(DMG), but is briefly turned ON in a period of time within oscillation time T_(OSC), to release some electric energy stored in capacitor CAC.

The flyback mode is a discontinuous conduction mode (DCM), because inductor current I_(M) flowing through primary winding LP stays at 0 A sometimes.

When operated in the flyback mode, if power controller 14 determines that compensation signal V_(COMP) exceeds a reference voltage V_(COMP-REF), power controller 14 exits the flyback mode and enters the ACF mode.

FIG. 4 enlarges the waveform of current-sense signal V_(CS) during low-side ON time T_(ON-L). When operated in the ACF mode, inductor current I_(M) through primary winding LP might be negative at the beginning of low-side ON time T_(ON-L), so current-sense signal V_(CS) is negative in that beginning. During low-side ON time T_(ON-L), as input voltage V_(IN) constantly increases the magnetic energy stored by primary winding LP, current-sense signal V_(CS) increases linearly over time until current-sense signal V_(CS) exceeds attenuated compensation signal V_(COMP-SC). Shown in FIG. 4, negative-current duration T_(ON-N) refers to the period of time when current-sense signal V_(CS) is negative, and positive-current duration T_(ON-P) the period of time when it is positive. Only if positive-current duration T_(ON-P) is longer than negative-current duration T_(ON-N), ACF power converter 10 is transferring and supplying energy to output voltage V_(OUT). From another perspective of view, if positive-current duration T_(ON-P) is very close to negative-current duration T_(ON-N) and the compensation signal V_(COMP) stays unchanged, it implies that load 13 is not heavy anymore, and should be a middle load or a light load.

It can be found from FIG. 4 that compensation signal V_(COMP) or attenuated compensation signal V_(COMP-SC), which are usually used to indicate the status of load 13, cannot represent the status of load 13 anymore, basically due to the existence of negative-current duration T_(ON-N). Therefore, it is a better choice to select positive-current duration T_(ON-P) and negative-current duration T_(ON-N) as indicators, instead of compensation signal V_(COMP), for determining whether to exit the ACF mode.

As shown in FIG. 2, in one embodiment of the invention, power controller 14 checks whether positive-current duration T_(ON-P) and negative-current duration T_(ON-N) have a predetermined relationship therebetween, to exit the ACF mode and enter the flyback mode. For example, when T_(ON-P)<T_(ON-N)+K_(T), power controller 14 exits the ACF mode and enters the flyback mode, where K_(T) is a positive constant. The predetermined relationship is not limited to the comparison between positive-current duration T_(ON-P) and negative-current duration T_(ON-N). In another embodiment of the invention, for example, power controller 14 checks energization duty cycle D_(ON-P), referring to T_(ON-P)/(T_(ON-p)+T_(ON-N)), to see if it is smaller than a predetermined value, so as to exit the ACF mode and enter the flyback mode.

In one embodiment of the invention, power controller 14 exits the ACF mode and enters the flyback mode right after the switching cycle in which positive-current duration T_(ON-P) and negative-current duration T_(ON-N) are found to have the predetermined relationship, but this invention is not limited to. In another embodiment of the invention, power controller 14 delays to exit the ACF mode and enter the flyback mode until positive-current duration T_(ON-P) and negative-current duration T_(ON-N) have continuously been found to have the predetermined relationship for a predetermined time period, 1 ms for example. This delay is especially beneficial during the test of load transient response. Supposedly this delay is 1 ms, and, under a test of load transient response, the status when load 13 is a light load does not last more than 1 ms before load 13 switches to become a heavy load. Under this test of load transient response, power controller 14 will continue to be operated in the ACF mode when load 13 briefly changes into a light load, and ACF power converter 10 expectedly has better transient response and more stable output voltage V_(OUT).

FIG. 5A shows the relationship between switching frequency f_(CYC) and compensation signal V_(COMP) for ACF power converter 10. When operated in the ACF mode and in the flyback back, the relationship is demonstrated by curves Cf_(CYC-ACF) and Cf_(CYC-FLY), respectively. Curve Cf_(CYC-ACF) clearly shows that switching frequency f_(CYC) is a constant f_(H) when operated in the ACF mode, and is independent from compensation signal V_(COMP). Curve Cf_(CYC-FLY) shows that when compensation signal V_(COMP) is between 4.3V and 0.7V switching frequency f_(CYC) and compensation signal V_(COMP) have a positive linear correlation with each other, meaning switching frequency f_(CYC) increases linearly as compensation signal V_(COMP) increases. In case the embodiment in FIG. 1 has a function of frequency jittering, curves Cf_(CYC-ACF) and Cf_(CYC-FLY) represent averages of switching frequency f_(CYC) when it is jittered during the ACF mode and the flyback mode respectively.

It is also shown in FIG. 5A that power controller 14 is operated in a burst mode when compensation signal V_(COMP) is around 0.5V, no matter which operation mode it was operated in previously. The burst mode can reduce the switching loss of high-side switch HSS and low-side switch LSS, and possibly increases the power conversion efficiency when supplying power to a light load or no load. If output current I_(O) is positive but very little, compensation signal V_(COMP) could go below 0.5V, causing power controller 14 to constantly turn OFF high-side switch HSS and low-side switch LSS and resulting in switching frequency f_(CYC) equal to 0, no power conversion at all. As power conversion pauses while output current I_(O) continues, output voltage V_(OUT) decreases and compensation signal V_(COMP) will go upward over time. Once compensation signal V_(COMP) exceeds 0.7V, power controller 14 resumes to operate in the flyback mode or the ACF mode that it was operated in before the power conversion paused, supplying power to output voltage V_(OUT). If output current I_(O) is still so little that the energy the ACF power converter 10 supplies to output voltage V_(OUT) exceeds the energy that load 13 consumes, output voltage V_(OUT) will go upward and compensation signal V_(COMP) eventually will go below 0.5V again, causing power conversion to pause once again. Therefore, if load 13 is always little, switching frequency f_(CYC) will alternate between being 0 Hz for a period of time and being non-zero Hz for another period of time. This kind of operation mode is known as a burst mode.

FIG. 5B demonstrates the relationship between signal peak V_(CS-PEAK) and compensation signal V_(COMP) regarding to the embodiment of FIG. 1. When operated in the ACF mode, signal peak V_(CS-PEAK) and compensation signal V_(COMP) have a relationship shown by curve CV_(CS-P-ACF); when operated in the flyback mode, they have a relationship shown by curve CV_(CS-P-FLY). Curve CV_(CS-P-ACF) indicates a positive, linear correlation between signal peak V_(CS-PEAK) and compensation signal V_(COMP), the higher compensation signal V_(COMP) the higher signal peak V_(CS-PEAK). Curve CV_(CS-P-FLY) indicates signal peak V_(CS-PEAK) as a constant V_(CS-REF) independent from compensation signal V_(COMP).

FIG. 5C demonstrates the graph of current I_(O) vs compensation signal V_(COMP) when the embodiment of FIG. 1 stays on a steady state, and the mode switching between the ACF mode and the flyback mode as well. When operated in the ACF mode, the relationship between output current I_(O) and compensation signal V_(COMP) is represented by curve CI_(O-ACF); when operated in the flyback mode, it is represented by curve CI_(O-FLY). It is supposed that output current I_(O) of ACF power converter 10 is initially below reference current I_(O-2), and, according to FIG. 5C, power controller 14 should be operated in the flyback mode. When output current I_(O) varies, compensation signal V_(COMP) changes accordingly, following curve CI_(O-FLY). In case that output current I_(O) steadily increases to exceed reference current I_(O-1), power controller 14 determines that compensation signal V_(COMP) is larger than reference voltage V_(COMP-REF), so it exits the flyback mode and enters the ACF mode. Once it enters the ACF mode, compensation signal V_(COMP) increases dramatically to approach to the value corresponding to reference current I_(O-1) on curve CI_(O-ACF). Now when output current I_(O) varies, compensation signal V_(COMP) changes accordingly, following curve CI_(O-ACF). In case that power controller 14 determines that positive-current duration T_(ON-P) and negative-current duration T_(ON-N) have reached the predetermined relationship, output current I_(O) is about reference current I_(O-2), and power controller 14 exits the ACF mode to enter the flyback mode. Due to the operation mode switching from the ACF mode to the flyback mode, compensation signal V_(COMP) decreases dramatically to approach to the value corresponding to reference current I_(O-2) on curve CI_(O-FLY).

FIG. 6 demonstrates consecutive switching cycles T_(CYC) when ACF power converter 10 in FIG. 1 is operated in the flyback mode. As shown in FIG. 6, low-side signal DRV_(LS) switches low-side switch LSS to continuously and periodically generate N switching cycles T_(CYC), where N is 8 for example, an integer bigger than 1.

The waveforms in FIG. 6 from top to bottom are clock signal CLK, high-side signal DRV_(HS), low-side signal DRV_(LS), current-sense signal V_(CS), terminal voltage V_(SW), blank signal S_(BLAN), and count CNT.

Blank signal S_(BLAN) generated internally in power controller 14 defines blanking time T_(BLAN), which represents the minimum cycle time of the present switching cycle T_(CYC). Only when blanking time T_(BLAN) elapses, the current switching cycle T_(CYC) can conclude and a next switching cycle T_(CYC) can start. Blanking time T_(BLAN) is determined by load 13 for example. In one embodiment, blanking time T_(BLAN) is generated in response to compensation signal V_(COMP), and the relationship of maximum frequency f_(BLAN), the reciprocal of blanking time T_(BLAN), versus compensation signal V_(COMP) could be represented by curve CF_(CYC-FLY) in FIG. 5A.

Power controller 14 could have a counter to record count CNT of the switching cycles T_(CYC). When it is determined that N switching cycles T_(CYC) have appeared, the counter is reset to make count CNT 1, as shown in FIG. 6, to restart count CNT.

FIG. 7 demonstrates control method 60 in use of power controller 14. When count CNT is smaller than N, meaning the current switching cycle must be one of the first N−1 switching cycles, step 62 of control method 60, which checks whether the count CNT is N, will generate a negative result, and control method 60 proceeds to steps for a normal flyback cycle. In other words, each of the first N−1 switching cycles is deemed as a normal flyback cycle. When the count CNT is N, meaning the current switching cycle must be the Nth switching cycle, step 62 of control method 60 will generate a positive result, and control method 60 proceeds to steps for a modified flyback cycle. The Nth switching cycle is deemed as a modified flyback cycle. The count CNT is reset to be 1 in the end of the Nth switching cycle.

FIGS. 6 and 7 show that only one of N consecutive switching cycles T_(CYC) is a modified flyback cycle, and the rest are normal flyback cycles, but the invention is not limited to. According to embodiments of the invention, several consecutive ones of N consecutive switching cycles are modified flyback cycles and the rest are normal flyback cycles.

Taking the demonstration in FIG. 6 as an example, a difference between a normal flyback cycle and a modified flyback cycle can be found by scrutinizing the waveform of high-side signal DRV_(HS). Within a normal flyback cycle, high-side signal DRV_(HS) is always “0” in logic, keeping high-side switch HSS constantly turned OFF. Nevertheless, within a modified flyback cycle, even though high-side signal DRV_(HS) stays most of time at “0” in logic, it becomes “1” in logic shortly about at the end of the modified flyback cycle, turning ON high-side switch HSS for a short period of time. Accordingly, a switching cycle T_(CYC) for a modified flyback cycle includes high-side ON time T_(ON-H), as shown by the Nth switching cycle in FIG. 6.

Since high-side switch HSS is always turned OFF within a normal flyback cycle, the energy that the leakage inductance of primary winding LP is energized during low-side ON time T_(ON-L) will accumulate on capacitor CAC, so voltage V_(CO) increases switching cycle by switching cycle. Each modified flyback cycle, due to the brief high-side ON time T_(ON-H), could release a portion of the energy to output voltage V_(OUT), to increase the conversion efficiency. At the same time, voltage V_(CP) could accordingly reduce, avoiding low-side switch LSS from being damaged by an over-high voltage V_(CP) that stresses low-side switch LSS when low-side switch LSS is turned OFF.

According to embodiments of the invention, an active-clamp circuit needs a bleeder resistor no more, because voltage V_(CP) could decrease within a modified flyback cycle, so power conversion could be improved and manufacturing cost reduced. As demonstrated by ACF power converter 10 in FIG. 1, active-clamp circuit ACC is a no-loss active-clamp circuit because it includes no bleeder resistor.

Referring to FIG. 7, for a normal flyback cycle, step 64 a, using low-side signal DRV_(HS), turns ON low-side switch LSS to generate low-side ON time T_(ON-L) while making signal peak V_(CS-PEAK) a constant. Most of the signal waveforms during the 1^(st) switching cycle T_(CYC) in FIG. 6, for example, are self-explanatory in light of FIG. 3B and the related teaching. Blanking time T_(BLAN) starts at the same time when low-side ON time T_(ON-L) starts, and has a length in response to load 13. For example, the lighter load 13 the longer blanking time T_(BLAN). Within the 1^(st) switching cycle T_(CYC), blanking time T_(BLAN) covers low-side ON time T_(ON-L), demagnetization time T_(DMG), and a portion of oscillation time T_(OSC). Terminal voltage V_(SW) oscillates during oscillation time T_(OSC) within the 1^(st) switching cycle T_(CYC), producing peaks PK₁, PK₂ and valleys VY₁, VY₂ and VY₃.

Step 66 a in FIG. 7 waits until the end of blanking time T_(BLAN). FIG. 6 illustrates in the 1^(st) switching cycle T_(CYC) that blanking time T_(BLAN) ends about after the occurrence of peak PK₂.

Step 68 in FIG. 7 flows step 66 a, detecting whether a valley of terminal voltage V_(SW) happens. Step 70, following when it is determined that a valley happens, increases count CNT by 1 and concludes the current normal flyback cycle. Within the 1^(st) switching cycle T_(CYC) in FIG. 1, for example, valley VY₃ appears at moment t_(DET), so clock signal CLK concludes the 1^(st) switching cycle T_(CYC), count CNT increases by 1, and the 2^(nd) switching cycle T_(CYC) starts.

Shown in FIG. 7, steps 64 b and 66 b for a modified flyback cycle are the same with steps 64 a and 66 a respectively, and are not detailed for brevity. The Nth switching cycle T_(CYC) shown in FIG. 6 is a modified flyback cycle, where blanking time T_(BLAN) covers low-side ON time T_(ON-L), demagnetization time T_(DMG), and portion of oscillation time T_(OSC). Terminal voltage V_(SW) oscillates during oscillation time T_(OSC) within the Nth switching cycle T_(CYC), producing peaks PK₁, PK₂, PK₃ and valleys VY₁, VY₂ and VY₃

Step 72 in FIG. 7 follows step 66 b, detecting whether a peak of terminal voltage V_(SW) happens. Step 74, following when it is determined that a peak happens, turns ON high-side switch HSS, starting high-side ON time T_(ON-H). As shown by the Nth switching cycle T_(CYC) in FIG. 6, peak PK₃ is the 1^(st) peak after the end of blanking time T_(BLAN), so high-side ON time T_(ON-H) starts at about the moment when peak PK₃ appears. During high-side ON time T_(ON-H), voltage V_(CP) at the joint between high-side switch HSS and capacitor CAC might decrease slightly because voltage V_(CP) energizes primary winding LP.

According to some embodiments of the invention, one modified flyback cycle has only one high-side ON time T_(ON-H), and it appears only after the end of blanking time T_(BLAN), as exemplified by FIG. 6. This invention is not limited to however. Some embodiments of the invention might have more than one high-side ON time T_(ON-H) within one modified flyback cycle.

The duration of high-side ON time T_(ON-H) in each modified flyback cycle might be a predetermined constant according to embodiments of the invention. But this invention is not limited to. Some embodiments of the invention may have the duration of high-side ON time T_(ON-H) determined in response to voltage V_(CP) at the joint between high-side switch HSS and capacitor CAC, while power controller 14 detects winding voltage V_(AUX) via feedback pin FB to indirectly detect voltage V_(CP). For example, if power controller 14, during high-side ON time T_(ON-H), finds voltage V_(CP) is below a reference value, then power controller 14 ends high-side ON time T_(ON-H) in a modified flyback cycle.

Step 76 in FIG. 7, following step 74 after the end of high-side ON time T_(ON-H), generates dead time TD_(F) and then makes low-side signal DRV_(LS) to turn into “1” in logic from “0” at the moment when terminal voltage V_(SW) is about 0V. In other words, step 76 makes low-side switch LSS perform ZVS. Step 78 follows step 76, concluding the Nth switching cycle T_(CYC), and resetting count CNT to be 1, so as to let the next switching cycle T_(CYC) start.

From the embodiment shown by FIGS. 6 and 7, ACF power converter 10 acts like a quasi-resonant power converter when it is operated in a flyback mode, because a normal flyback cycle and a modified flyback cycle each ends at about the moment when a valley of terminal voltage V_(SW) appears, performing valley switching that is capable of reducing switching loss. This invention is not limited to however. It is not necessary for ACF power converter 10 to perform valley switching when operated in a flyback mode. For example, some embodiments of the invention might skip step 68 in FIG. 7, and start a next switching cycle right after the end of blanking time T_(BLAN).

Even though FIGS. 6 and 7 show that high-side ON time T_(ON-H) in a modified flyback cycle starts at about the moment when a peak appears, but this invention is not limited to. Some embodiments of the invention might have step 72 in FIG. 7 skipped or modified. Some embodiments of the invention have step 72 modified to detect a next valley after the end of blanking time T_(BLAN) and start high-side ON time T_(ON-H) at about the moment when the next valley appears, for example. Other embodiments of the invention nevertheless have step 72 skipped, to start high-side ON time T_(ON-H) right after the end of blanking time T_(BLAN).

N is a constant integer according to embodiments of the invention, but this invention is not limited to. N might be adaptively changed in some embodiments of the invention. For example, power controller 14 could detect voltage V_(CP), via the help of feedback pin FB and auxiliary winding LA, during high-side ON time T_(ON-H). Voltage V_(CP) is the voltage at an end of primary winding LP when high-side switch HSS is turned ON. If voltage V_(CP) is higher than a top boundary of a predetermined acceptable range, N seems too large and is going to decrease by 1 at the end of the Nth switching cycle, implying the increased frequency for a modified flyback cycle to appear. On the other hand, if voltage V_(CP) is lower than a bottom boundary of the predetermined acceptable range, N seems too small and is going to increase by 1 at the end of the Nth switching cycle. Accordingly, voltage V_(CP) is adaptively controlled to substantially stay within the acceptable range.

While the invention has been described by way of example and in terms of preferred embodiment, it is to be understood that the invention is not limited thereto. To the contrary, it is intended to cover various modifications and similar arrangements (as would be apparent to those skilled in the art). Therefore, the scope of the appended claims should be accorded the broadest interpretation so as to encompass all such modifications and similar arrangements. 

What is claimed is:
 1. A control method in use of a power converter using an active-clamp flyback topology, the power converter comprising an active-clamp circuit connected in parallel with a primary winding of a transformer, the active-clamp circuit comprising a high-side switch and a capacitor connected in series, the power converter further comprising a low-side switch connecting the primary winding to a first power line, the control method comprising: switching the low-side switch to generate consecutive switching cycles including a modified flyback cycle and a normal flyback cycle; making each of the consecutive switching cycles having a cycle time not less than a blanking time generated in response to a load of the power converter; constantly turning the high-side switch OFF during the normal flyback cycle; and turning the high-side switch ON after the blanking time during the modified flyback cycle to perform zero-voltage switching for the low-side switch.
 2. The control method as claimed in claim 1, wherein a terminal voltage at the low-side switch oscillates after a demagnetization time to generate at least one peak and one valley, the control method comprising: detecting whether the peak appears; and turning the high-side switch ON about when the peak appears.
 3. The control method as claimed in claim 1, wherein a terminal voltage at the low-side switch oscillates after a demagnetization time to generate at least one peak and one valley, the control method comprising: detecting whether the valley appears; and turning the high-side switch ON about when the valley appears.
 4. The control method as claimed in claim 1, wherein only one of the consecutive switching cycles is a modified flyback cycle.
 5. The control method as claimed in claim 1, comprising: switching the low-side switch to repeatedly generate the consecutive switching cycles.
 6. The control method as claimed in claim 5, wherein the consecutive switching cycles are N in total and N is a predetermined constant.
 7. The control method as claimed in claim 5, comprising: providing a counter counting a number of the consecutive switching cycles.
 8. The control method as claimed in claim 1, wherein the consecutive switching cycles are N in total and the transformer comprises an auxiliary winding, the control method comprising: adaptively changing N to keep a voltage at an end of the primary winding substantially staying within an acceptable range.
 9. The control method as claimed in claim 1, comprising: during the modified flyback cycle, turning the high-side switch ON only after the blanking time.
 10. A power converter using an active-clamp flyback topology, comprising: a transformer with a primary winding; a low-side switch for connecting the primary winding to a first power line; a high-side switch connected in series with a capacitor to form an active-clamp circuit connected in parallel with the primary winding; and a control circuit, configured to control the high-side switch and the low-side switch in response to a compensation signal and a current-sense signal; wherein the control circuit selectively operates the power converter in one of operation modes including a flyback mode; when operated in the flyback mode, the control circuit switches the low-side switch to generate consecutive switching cycles including a normal flyback cycle and a modified flyback cycle; during a normal flyback cycle, the control circuit constantly turns the high-side switch OFF; during a modified flyback cycle, the control circuit turns ON the high-side switch after a blanking time to perform zero-voltage switching for the low-side switch; and the control circuit determines the blanking time in response to the compensation signal.
 11. The power converter as claimed in claim 10, wherein when operated in the flyback mode, the control circuit makes the current-sense signal have a signal peak independent from the compensation signal.
 12. The power converter as claimed in claim 10, wherein the consecutive switching cycles are N in total and N is a predetermined constant.
 13. The power converter as claimed in claim 10, wherein the consecutive switching cycles are N in total and N is determined in response to a voltage at an end of the primary winding.
 14. The power converter as claimed in claim 10, wherein a terminal voltage at the low-side switch oscillates after a demagnetization time to generate at least one peak and one valley, the control circuit detects whether the peak appears, and the control circuit turns the high-side switch ON about when the peak appears.
 15. The power converter as claimed in claim 10, wherein a terminal voltage at the low-side switch oscillates after a demagnetization time to generate at least one peak and one valley, the control circuit detects whether the valley appears, and the control circuit turns the high-side switch ON about when the valley appears.
 16. The power converter as claimed in claim 10, wherein the control circuit turns ON the high-side switch right after an end of the blanking time. 